Radio frequency receiving circuit having a passive monopulse comparator

ABSTRACT

A passive MMIC monopulse comparator includes a plurality of lumped element hybrids having pi and T filter structures. In one embodiment, a first hybrid receives first and second downcoverted signals and provides a first output signal to a third hybrid and second output signal to a fourth hybrid. A second hybrid receives third and fourth downconverted signals and provides respective third and fourth output signals to the third and fourth hybrids. The third hybrid provides sum and elevation channel signals and the fourth hybrid provides azimuth and G channel signals.

RELATED APPLICATIONS

This application is a continuation-in-part of U.S. patent applicationSer. No. 09/100,689, filed on Jun. 19, 1998 now U.S. Pat. No. 6,100,841.

GOVERNMENT RIGHTS

Not applicable.

FIELD OF THE INVENTION

This invention relates generally to receiver circuits and moreparticularly to radio frequency (RF) receiver circuits.

BACKGROUND OF THE INVENTION

As is known in the art, a radar system generally includes an antenna, atransmitter and a receiver. In general overview, the transmittergenerates an electromagnetic signal which is emitted or radiated throughthe antenna. The radiated electromagnetic signal propagates in apredetermined region of space and intercepts one or more objects in thepath of the electromagnetic radiation. Portions of the electromagneticradiation reflect off the objects and propagate back towards the radarsystem where the reflected signals are detected by the receiver. Suchreflected signals are sometimes referred to as return or echo signals.

If the radar system employs a directive antenna, a relatively narrowbeam of electromagnetic radiation is emitted and the direction fromwhich the return signals propagate and hence the bearing of the objectmay be estimated. The distance or range to the reflecting object can beestimated by transmitting signal pulses and measuring the time periodbetween the transmission of the transmitted pulse and reception of thereturn signal pulse.

One particular type of radar system is a monopulse radar system. Amonopulse radar system refers to a radar system which obtains a completemeasurement of an object's angular position by transmitting a singlesignal pulse and receiving the corresponding return or echo pulse.Together with a range measurement performed with the same pulse, theobject position in three dimensions is determined completely. Typically,a series or train of echo pulses is employed to make a large number ofrepeated measurements and produce a refined estimate of the object'sposition.

A monopulse receiving system typically includes a monopulse circuitwhich receives signals from the antenna and forms sum and differencemonopulse output signals. The sum and difference signals are formed bycombining received antenna signals in a particular manner. The signalscan be combined using circuits referred to as hybrid circuits. Thehybrid circuits may be provided as so-called magic-T or rat racecircuits which receive signals fed thereto and add and/or subtract thesignals in a known manner. Such hybrid circuits can be fabricated usingeither printed circuit or waveguide transmission lines.

To determine the location of an object in a single angular coordinate(e.g. either azimuth or elevation), the monopulse circuit need onlyinclude a single hybrid circuit and thus the monopulse circuit isrelatively compact. To determine the location of an object in twoangular coordinates (e.g. both azimuth and elevation), the monopulsecircuit requires multiple hybrid circuits. Thus, conventional monopulsecircuits capable of determining the location of an object in two angularcoordinates can become relatively large.

The monopulse sum and difference signals can be formed either at thetransmitted signal frequency or, after down conversion of a returnsignal, at a lower frequency. The transmit signal frequency is typicallyin the microwave or millimeter wave frequency range. When the monopulsesum and difference signals are formed at the transmitted signalfrequency, the monopulse is typically coupled directly to the antennawith relatively few, if any, circuits disposed between the antennaoutput ports and the monopulse input ports. The operations to generatemonopulse sum and difference signals typically are performed atmicrowave or millimeter wave frequencies by the hybrid circuits whichare typically fabricated using either printed circuit or waveguidetransmission lines.

Obtaining the sum and difference signals at the transmitted signalfrequency (i.e., before any down conversion) reduces the amount ofadditional errors which may be otherwise introduced into the signalsused to form the monopulse signals by circuits (e.g. mixer circuits)coupled between the antenna output ports and the monopulse input ports.For example, to form the monopulse signals after down conversion of areturn signal to a lower frequency it is necessary to couple a mixer orother frequency translation device between the antenna output ports andthe monopulse input ports. Practical frequency translation devices (e.g.mixer circuits) introduce errors into the signals which are combined inthe monopulse circuit to provide the monopulse output signals.

Typically, a single sum channel and a pair of difference channels areformed by the monopulse circuit to allow resolution of two angularcoordinates. In systems which utilize a conventional waveguide multimodehorn feed, a waveguide monopulse network can process a radar returnsignal to generate monopulse sum and difference signals which propagatein appropriate monopulse sum and difference channels. The radiofrequency (RF) signals propagating through the monopulse channels areconverted to intermediate frequency (IF) signals using waveguide mixers.The IF signals are fed to an IF receiver for additional processing.

One problem with this RF waveguide approach to implementing themonopulse network is that the monopulse circuit is relatively large andmust be fabricated using relatively expensive and time consumingprecision machining or electroforming techniques. This is particularlytrue in those system which operate in the millimeterwave frequencyrange. To overcome this drawback, systems operating at millimeterwavelength,frequencies can downconvert received signals to anintermediate frequency prior to monopulse processing. With thisapproach, monopulse processing may be performed at the intermediatefrequency in lieu of monopulse processing performed at the higherfundamental or transmit frequency. While the circuit fabricationtolerances are generally less severe at lower frequencies, there is aconcomitant increase in the size of waveguide circuit components. Thus,the use of waveguide transmission lines to process and convert themonopulse information (especially at millimeter wave frequencies) is nota practical low cost solution suitable for high volume production.

To further complicate matters, projectiles such as missiles andsubmunitions having a relatively small diameter require relatively highresolution monopulse receivers to enable accurate tracking of a target.Conventional monopulse receiving systems operating in the 1 gigahertz(GHz) to 20 GHz frequency range do not provide the angular resolutionneeded to accurately track targets. Furthermore, the size of RF circuitcomponents which operate in the 1 GHz to 20 GHz range are physically toolarge and cumbersome to be packaged in many small projectiles.Therefore, operation at millimeter wave frequencies above 30 GHz isrequired.

Missile seeker systems having a relatively large diameter typicallyoperate at microwave frequencies and form monopulse output receivesignals with comparator networks provided from hybrid circuit componentsimplemented using stripline, coaxial or waveguide transmission media.The monopulse output signals are typically fed to amplifiers having arelatively high gain and a relatively low noise figure. The amplifiedsignals are subsequently downconverted to an appropriate intermediatefrequency (IF) by a radio frequency (RF) microwave mixer module. Forthose applications in which the monopulse receiver must be disposed in aprojectile having a relatively small diameter, however, the signaltransmission losses and overall size of conventional receiver systemsadversely impact seeker performance. Operation at higher frequenciessuch as millimeter wave (MMW) frequencies is a necessity to achieve therequisite resolution but there are limitations in the availability ofreceiver devices which operate at such frequency bands. For example, itis relatively difficult and expensive to provide RF devices having theperformance characteristics (e.g., noise figure, power handling, powerlimiting, etc.) required for efficient active seeker operation in theMMW frequency range.

The complexity of radar systems operating in the millimeter wavefrequency band will be appreciated when it is recognized that at anoperating frequency of 94 GHz, for example, dimensions of a conventionalrectangular waveguide are in the order of 0.050 to 0.100 inches, withtolerances of better than 0.001 inches required in many criticalassemblies. Although it may be possible to fabricate suchmillimeter-wave waveguide structures at somewhat reduced cost usingmodem fabrication techniques, the expense associated with tuning andtesting such critically toleranced hardware is often cost prohibitive.

Furthermore, the problems of packaging and tuning a millimeter-waveseeker in a conventional submunition will be appreciated when it isrecognized that a monopulse seeker with a monopulse tracking capabilityutilizing waveguide components may well require in excess of twentydifferent waveguide components to control the routing and duplexing ofthe various signals coming from the transmitter and returning to thereceivers. If a monopulse capability were required, then all of theforegoing waveguide components would be required to track from channelto channel in both amplitude and phase.

At an operating frequency of 94 GHz, each one thousandth of an inch inthe length of a waveguide transmission line is equivalent to about 2° ofphase. It should, therefore, be appreciated that it is relativelydifficult to obtain inexpensively the requisite phase and amplitudetracking between various receiver channels.

Another problem inherent in millimeter-wave radar seekers utilizingwaveguide devices is that of providing sufficient isolation between atransmitter and receiver. This problem is exacerbated by the fact thatwaveguide switches and circulators which can withstand relatively highpower transmit signals and provide a high degree of isolation are notgenerally available in a compatible size at relatively high operatingfrequencies.

It would, therefore, be desirable to provide a relatively compactmonopulse receiver having a relatively low noise figure which operatesin the millimeter wave frequency range and which can operate in a systemwhich includes a transmitter which transmits signals having relativelyhigh power levels.

SUMMARY OF THE INVENTION

In accordance with the present invention, a radio frequency (RF) systemincludes an antenna having a plurality of antenna ports and a pluralityof protection circuits each of the protection circuits having a firstport coupled directly to a respective one of the plurality of antennaports and a second port. In response to a first control signal, eachprotection circuit allows signals to propagate from a respective one ofthe antenna ports to the second port of the respective protectioncircuit along a signal path having a relatively low insertion losscharacteristic. Each protection circuit is also responsive to a secondcontrol signal in a first direction between the first protection circuitport and the second protection circuit port and responsive to a secondcontrol signal which isolates the first protection circuit port from thesecond protection circuit port. With this particular arrangement, acompact RF system is provided. By coupling the antenna ports directly tothe ports of the protection circuit, the RF system can operate in areceive mode and be protected from transmit signal having high signallevels generated by a transmitter circuit during a transmit operatingmode. To operate in a receive mode, the protection circuit is biased toprovide a signal path having a relatively low insertion losscharacteristic to signals propagating from the antenna ports through theprotection circuit ports. During the transmit mode, the protectioncircuit is biased to provide a signal path having high insertion losscharacteristic from signals propagating from the antenna ports. In oneparticular embodiment, the RF system further includes a plurality ofmixers, each having a first port coupled to a respective one of theplurality of protection circuit ports and a second port for receiving amixer bias signal and a third port for providing a frequency shiftedsignal. By coupling the mixers to the protection circuit ports, acompact receiver assembly is provided. An amplifier can be coupled tothe third port of each mixer to thus provide a system having arelatively high gain characteristic while also providing a system whichprovides a relatively low noise figure. Also a monopulse can be coupledto the output ports of the amplifiers to provide an RF monopulsereceiving system. In a preferred embodiment, the mixers, amplifiers andmonopulse circuit are provided as monolithic microwave integratedcircuits (MMICs) and thus the RF system utilizes a relatively smallphysical area. Furthermore the antenna can be provided as a corrugatedhorn having a moding structure disposed in a base portion thereof tocouple signals between the antenna input and the antenna ports in thebase structure of the corrugated horn which are coupled to the firstport of each of the protection circuits. The receiving system may alsoinclude a calibration signal injection circuit coupled to the protectioncircuit to inject a calibration signal into the receiving system. In oneparticular embodiment, the protection circuit is a latching ferriteisolator matrix which includes a plurality of isolators each havingfirst, second and third ports with the calibration signal injectioncircuit coupled to the third port of each of the plurality of isolators.

In accordance with a further aspect of the present invention, a radiofrequency (RF) monopulse receiver includes a plurality of mixers, eachhaving an RF signal port, a local oscillator (LO) signal port and anintermediate frequency (IF) signal port and each of the mixerscomprising one or more mixer diode anti-parallel pairs and means forcoupling RF energy to the RF signal port of each of the plurality ofmixers. The RF monopulse receiver further includes a plurality of IFamplifiers, each of the IF amplifiers having an amplifier input portcoupled to the IF signal port of a respective one of the plurality ofmixers and an amplifier output port coupled to a respective one of aplurality of input ports of a monopulse comparator network. In responseto appropriate input signals fed thereto, the monopulse comparatornetwork provides monopulse output signals at output ports thereof. Withthis particular technique, a compact millimeter wave monopulse receiverhaving a relatively low noise figure is provided. By arranging alatching ferrite isolator matrix protection circuit between an antennaand the mixer ports, the RF monopulse receiver is protected from highpower transmit signals. Furthermore the latching ferrite isolator matrixallows use of a receiver circuit architecture which allows the compactmillimeter wave monopulse receiver circuit to operate in an RF radarsystem having a relatively high transmit power. In one embodiment, theRF monopulse receiver is suitable for use in an active missile seekersystem for example. The RF receiver can be used directly in a smallsubmunition or alternatively, can function as the monopulse receiver fora higher resolution quasi-optically fed, antenna having an aperture muchlarger than the diameter of the horn antenna. In one embodiment, acryogenic cooling system is coupled to the receiver to provide areceiver noise figure which is lower than the noise figure achieved whenthe receiver operates at ambient temperatures.

In accordance with a still further aspect of the present invention, anRF monopulse receiver includes a circuit assembly having a plurality ofRF input ports and a plurality of IF output ports. The circuit assemblyincludes (a) a housing, (b) a plurality of subharmonically pumped mixercircuits disposed in the housing, each of the mixer circuits having anRF signal port, an LO signal port and an IF signal port and each of themixer circuits including: (1) a plurality of mixer diode substratesdisposed in the housing, each of the mixer diode substrates having adiode mounting region and a transmission coupling region which projectsinto an RF feed region which is formed by providing an opening in ahousing cover disposed over the diode mounting region of the pluralityof substrates; (2) an antiparallel diode pair disposed on the diodemounting region of each of the plurality of mixer diode substrates; (3)an LO distribution circuit coupled between the LO signal port of eachsubharmonically pumped mixer circuit and an LO signal source; (4) an IFdistribution circuit coupled between the IF signal port of eachsubharmonically pumped mixer circuit and an IF output port of the RFmonopulse receiver; (c) an RF feed circuit, coupled to the housing, (d)means for coupling RF energy to the RF signal port of each of theplurality of subharmonically pumped mixer circuits; (e) a monopulsesubstrate; and (f) a monolithic microwave integrated circuit (MMIC)monopulse comparator network disposed on the monopulse substrate, theMMIC monopulse comparator network having a plurality of monopulsecircuit input ports, each of the plurality of monopulse circuit inputports coupled to a respective one of the IF ports of the plurality ofsubharmonically pumped mixer circuits and having a plurality ofmonopulse circuit output ports coupled to the IF output ports of themonopulse substrate. With this particular arrangement, an RF monopulsereceiver suitable for use in the W-band frequency range is provided. Thesystem can further include a corrugated horn antenna having a modingstructure in a base portion thereof to provide four separate antennaports. The antenna base ports are coupled to the RF ports of the mixercircuits providing signals between the antenna ports in the basestructure of the corrugated horn and the RF port of the mixer circuits.A protection circuit can be included in a waveguide signal path disposedbetween the antenna ports and the RF input port of the mixer circuits.In one particular embodiment, the protection circuit is provided as alatching ferrite isolator matrix which includes a plurality of isolatorseach having first, second and third ports. The RF monopulse receiver canalso include a calibration signal injection circuit coupled to theprotection circuit to inject a calibration signal into the receiver.When the protection circuit is provided as the latching ferrite isolatormatrix, a calibration signal injection circuit can be coupled to thethird port of each of the plurality of isolators. In response to a firstcontrol signal, the latching ferrite isolators allow signals topropagate in a first direction between a first pair of isolator portsand in response to a second control signal, the latching ferriteisolators allow signals to propagate in a second direction between asecond pair of isolator ports. With such an arrangement, the RFmonopulse receiver can operate without being damaged in those RF systemswhich include a transmitter. Furthermore, the housing may be provided asa single housing having both RF and IF circuit components disposedtherein including the MMIC monopulse comparator network. Alternatively,the housing can be provided from an RF housing and an IF housing whichare physically and electrically coupled together. The RF signalcomponents including the RF mixer and an amplifier are disposed in theRF housing and the MMIC monopulse comparator network and an amplitudeadjustment circuit and phase shifter are disposed in the IF housing witha plurality of RF interconnect signal paths providing RF signal pathsbetween the RF housing and the IF housing. The use of MMIC LNA's and anovel MMIC monopulse along with the development of a low conversionloss, low noise figure, W-Band mixer leads to a relatively small andefficient W-Band monopulse receiver which is compatible with smalldiameter missiles and submunitions and provides enhanced sensitivity tothus allow the seeker to accurately track targets. This enhancedsensitivity is achieved with the inclusion of cooling hardware which isused to locally cool the mixer diodes.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing features of the invention, as well as the invention itselfmay be more fully understood from the following detailed description ofthe drawings, in which:

FIG. 1 is a schematic diagram of a receiving system;

FIG. 1A is a schematic diagram of a receiving system having acalibration system coupled thereto;

FIG. 2 is an exploded isometric view of a receiving system;

FIG. 2A is a side view of a multimode corrugated feed horn;

FIG. 2B is an end view of the multimode feed horn;

FIG. 2C is a plan view of a receiver;

FIG. 2D is a plan view of an RF mixer circuit taken along lines 2D—2D ofFIG. 2C;

FIG. 3 is a plan view of a receiver;

FIG. 4 is an exploded isometric view of a portion of a receiving systemhaving a cooling system;

FIG. 5 is a schematic representation of a passive monopulse comparatorin accordance with the present invention;

FIG. 6 is a schematic representation of a lumped element hybrid thatforms a part of the passive monopulse comparator of FIG. 5;

FIG. 7 is a schematic representation showing further details of thelumped element hybrid of FIG. 6;

FIG. 8 is a circuit diagram of the monopulse comparator of FIG. 5; and

FIG. 9 is a graphical depiction of signal insertion loss versusfrequency for a device fabricated in accordance with the presentinvention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring now to FIG. 1, a radio frequency (RF) receiving system 10includes an antenna 12 having a plurality of antenna output ports 12a-12 d. The antenna 12 may be provided as any type of antenna capable ofreceiving signals in a desired frequency range. The antenna 12 may beprovided, for example, as a horn antenna, an array antenna or any othertype of antenna capable of receiving RF signals at a first end 13 andproviding output signals at the antenna ports 12 a-12 d.

The antenna ports 12 a-12 d are coupled to respective input ports of aprotection circuit 16. In one particular embodiment, the antenna 12 isprovided as a corrugated waveguide horn responsive to signals in theW-band frequency range and having a moding structure 14 provided in abase portion thereof to provide separate antenna ports. The antenna 12receives RF energy at the first end 13 and the moding structure 14separates RF energy propagating in the horn 12 for transmission alongfour separate transmission line signal paths. In one embodiment, thetransmission lines are provided as waveguide transmission lines having asize and shape selected to facilitate efficient propagation of RFsignals in the frequency range of interest.

Protection circuit 16 can prevent signals having relatively highamplitude levels from propagating to the remaining circuit components ofsystem 10 thereby reducing the possibility of such components beingdamaged by signals having excessive signal amplitudes.

Protection circuit 16 is provided, in one embodiment, as a latchingferrite isolator matrix, which includes a plurality of latching ferriteisolators 18 a-18 d generally denoted 18. Each of the isolators 18 hasfirst, second and third ports 19 a-19 c with a termination 22 coupled toports 19 c. In response to a first control signal provided to isolators18 a-18 d through a bias line 20, the isolators 18 are polarized orbiased such that signals propagate through isolators 18 in a firstdirection. For example, in response to a first control signal providedto isolators 18, RF signals provided to port 19 a are coupled in aclockwise direction to port 19 b and are isolated from port 19 c.

In response to a second control signal provided to isolators 18, theisolators 18 are biased such that signals propagate through isolators 18in a second different direction. For example, signals provided to port19 c are coupled in a counterclockwise direction through the isolator 18to port 19 b and are isolated from port 19 a. The first and secondcontrol signals may be provided, for example, as current signals havinga relatively high amplitude over a relatively short duration of time(i.e. “current spikes”) which bias the isolators 18 in first and secondopposite directions.

With the protection circuit 16 biased to allow signals to propagate fromrespective ones of the antenna ports 12 a-12 d through port 19 a to port19 b of the respective isolators 18 a-18 d, the signals propagate fromthe antenna 12 through the protection circuit 16 to a receiver 23 atinput ports 23 a-23 d. In this particular embodiment, receiver 23 ishere shown as a monopulse receiver.

The receiver 23 includes an RF circuit module 24 having four likereceiver channels 25 a-25 d. Taking receiver channel 25 a asrepresentative of the receiver channels 25 b-25 d, receiver channel 25 aincludes a transition circuit 26 having an input port 26 a coupled toreceiver input port 23 a and an output port 26 b coupled to an RF port28 a of an RF mixer 28. A mixer bias signal is fed through an RF circuitmodule port 27 a to a mixer bias port 28 b. In this particularembodiment, the mixer bias signal is provided as a local oscillator (LO)bias signal. In an alternate embodiment, however, the mixer can bebiased with a DC bias signal which can be fed to mixer separately fromthe LO signal. The LO signal and RF signals fed to ports 28 a, 28 brespectively, are combined as is known to provide an intermediatefrequency (IF) signal at a third mixer port 28 c.

The mixer IF port 28 c is coupled to an input port of an IF amplifier 30which receives the IF signal fed thereto and provides an amplifiedoutput signal at an output port thereof. Thus each of the receiverchannels 25 a-25 d provide IF output signals at output ports thereof.

The IF output signals are coupled to respective ones of input ports 34a-34 d of an IF module 34. The IF module 34 receives the IF signals fedthereto and provides each of the IF signals to respective amplitude andphase adjustment circuits 36. In one particular embodiment, the phaseand amplitude adjustment circuit includes a variable attenuator 37 and asix-bit phase shifter 38 with which the amplitude and phase adjustment,respectively, are accomplished. Those of ordinary skill in the art willappreciate that the phase shifter 38 may be provided having fewer orgreater than six bits. The number of bits included in the phase shifter38 is selected in accordance with a variety of factors including but notlimited to the desired accuracy of the corrected signal, the number ofcontrol lines required to operate the phase shifter, the package sizeand weight, etc. With a six-bit phase shifter the least significant bithas a value of 5 degrees.

The appropriately amplitude and phase adjusted signals are coupled fromeach of the phase and amplitude adjustment circuits 36 to a monopulsecircuit 40 at input ports 40 a-40 d. The monopulse circuit 40 combinessignals fed to the input ports 40 a-40 d thereof and provides monopulseoutput signals (Σ, ΔEL, ΔAZ, Q) at ports 41 a-41 d.

It should be noted that in a preferred embodiment, monopulse 40 may beprovided as a passive monolithic microwave integrated circuit (MMIC)monopulse circuit. In alternate embodiments, however, it may bedesirable to provide the monopulse circuit 40 as an active MMIC circuit.Alternatively still, the monopulse circuit may be constructed from atransmission media other than a media which is suitable for fabricationof a MMIC circuit.

Referring briefly to FIG. 1A, in an alternate embodiment, the system 10′includes a calibration circuit 44 which includes a calibration signalsource 45 which provides a calibration signal through a protectioncircuit 46 which may be provided as an isolator or switch, for example.In a preferred embodiment, the calibration signal provided by thecalibration signal source 45 corresponds to a substantially puresinusoidal signal having a predetermined signal amplitude.

The calibration signal is coupled through protection circuit 46 to aninput port 47 of a power divider 48. In this particular embodiment,power divider 48 includes output ports 48 a through 48 d each of whichare coupled to a respective one of the isolators 18 a-18 d at isolatorports 19 c.

The power divider 48 is here provided having a number of output portscorresponding to the number of receiver channels. However, in someapplications, it may not be necessary to provide a calibration signal toeach of the receiver channels. Thus, generally, the power divider 48provides a calibration signal for each receiver channel 25 to becalibrated.

In some applications, it may be desirable to remove the terminations 22(FIG. 1) coupled to the ports 19 c of the isolators 18 to allow acalibration signal to be fed thereto while in other applications it maybe desirable to couple the calibration signals to the isolator ports 19c through a switch (not shown). In this approach the common port of theswitch would be coupled to isolator port 19 c, a first switch arm wouldbe coupled to a termination and a second switch arm would be coupled toreceive a calibration signal from the calibration circuit 44. In stillother applications it may be desirable to couple the calibration signalto a portion of the receiver channels.

The receiving system 10 operates in either a calibration mode or areceiving mode. To place the receiving system in the calibration mode,the isolators 18 are biased such that signals propagate in acounterclockwise direction therethrough such that signals propagate fromport 19 c to port 19 b. The protection circuit 46 provides a relativelylow insertion loss signal path between the signal source 45 and thepower divider circuit 48.

In the calibration mode, the calibration signal source 45 providescalibration signals to port 19 c of each isolator 18. The calibrationsignals are coupled from isolator ports 19 c to ports 19 b and providedas input signals to the RF circuit module 24 at ports 23 a-23 d. The RFcircuit module 24 provides in response to the calibration signalfrequency translated (i.e. a downconverted) calibration signals at theoutput ports 24 a-24 d.

The IF module 34 receives the downconverted calibration signals from RFmodules ports 24 a-24 d and provides calibration monopulse outputsignals at ports 41 a-41 d. The amplitude and phase of the calibrationoutput signals outputs 41 a-41 d can be monitored and correlated withexpected channel-to-channel signal characteristics to provide errorcorrection for the measured output signals. The expectedchannel-to-channel signal characteristics are defined during initialreceiver integration and alignment based on known operating conditionswhich yield acceptable monopulse performance. Thus, the calibration modeallows correction of channel-to-channel errors resulting from theoperational environments which cannot be predicted in advance such as(non-uniform module heating).

A second calibration circuit for introducing a calibration signal to IFmodule 34 includes an IF calibration signal port 42 through which acalibration signal is fed to a coupler circuit 43. The coupler circuit43 couples a predetermined portion of the calibration signal to theinput of the amplitude and phase adjustment circuit 36. The calibrationsignal propagates through circuit 36 to the monopulse 40 and calibrationoutput signals are provided at ports 41 a-41 d. Here, the IF calibrationsignal is shown as being fed to only one channel of IF module 34. Itshould be appreciated, however, that the IF calibration signal could befed to any single channel of IF module 34 or alternatively each channelof IF module could include a coupler 43 and the IF calibration signalcould be fed (simultaneously or not) to all channels of IF module 34.

Referring again to FIG. 1, it should also be noted, that in someapplications, it may be desirable to utilize the receiving system 10 asa feed circuit for a reflector antenna. This is accomplished by aligningthe multimode feed horn 12 and the focal plane of a reflector. In otherapplications, however, the multimode feed horn 12 may be used directlyas a monopulse antenna.

A optional cooling system 49 may be coupled to active circuit componentssuch as mixers 28 and amplifiers 30 in each receiver channel 25 to coolactive devices such as the mixer diodes in the components 28, and fieldeffect transistors (FETs) in the amplifiers 30 to thus lower the noisefigure of the receiver 23. The cooling system 49 may be used to cooleither or both of mixers 28 and amplifiers 30 respectively.

To provide the system 10 having a compact size, the amplifiers 30,amplitude and phase adjustment circuits 36, phase shifters 38 andmonopulse circuit 40, can all be provided as monolithic microwaveintegrated circuits (MMICs). Such an assembly results in a receiver 23having a relatively low noise figure at room temperatures and having asize which allows use of the receiving system 10 in missiles, munitionsand other projectiles having relatively small diameters.

Referring now to FIG. 2, a millimeter wave monopulse receiver 50includes a corrugated horn assembly 51 having a first aperture with apressure window 52 disposed therein. The pressure window 52 provides asealing mechanism for the receiver 50 for reasons which will beexplained more fully hereinbelow. A second end of the corrugated horn 51includes a base plate 53 having a plurality of mounting holes 54provided therein.

Corrugated horn 51 and pressure window 52 provide a relatively lowinsertion loss characteristic to signals having frequencies within apredetermined frequency band while still providing a sealed aperture viapressure window 52. In one particular embodiment, the corrugated horn 51is suitable for coupling efficiently to a W-Band Quasi-optical GuassianBeam input signal.

As may be more clearly seen in FIGS. 2A, 2B, also formed in thetransition region between the horn flare and the baseplate is a modingstructure which separates the captured energy into four identicalwaveguide transmission lines 57. Referring briefly to FIG. 2B waveguides57 formed in the flange 53 can be clearly seen. Projecting from asurface of flange 53 are optional alignment pins 54 a which engages acorresponding hole in the surface of a switch assembly 55 (FIG. 2) tothus align waveguide apertures 57 in the flange 53 with like-shapedapertures 57 in the switch assembly 55.

Referring again to FIG. 2, the waveguide energy passes through thelatching ferrite isolator (switch) matrix 55 which is primarily used forprotection purposes. The switch matrix 55 also allows for the injectionof a W-Band calibration tone by feeding a calibration signal having afrequency in the W-band frequency range through one or more signalport(s) of the isolator(s) corresponding to isolator port 19 c of FIG.1.

In a receive mode of operation, the horn 51 receives RF signals andcouples the received RF energy via a moding structure into the fourrectangular shaped waveguide transmission lines 57 disposed in theflange 53 at the second or base end of the corrugated horn 51. Flange 53is disposed over a first surface of the switch assembly 55 such thatalignment holes and pin 54, 54 a align with corresponding holes 59 inthe switch assembly 55.

The switch assembly 55 may be a Latching Ferrite Isolator Matrix of thetype provided by Electromagnetic Systems (EMS) Corporation andidentified as model no. 449D-68. When receiver 50 is disposed in amissile or other projectile which includes a transmitter, the switchassembly 55 prevents transmitter leakage signals impinging on the horn51 from propagating to a receiver circuit 56 since a relatively highpower leakage signal could damage circuit components of the receivercircuit 56.

Since the time during which an RF transmit signal pulse could leak intothe antenna 51 is known, the switch assembly 55 can be actively switchedbetween transmit and receive states at appropriate points in time. Whenthe switch assembly 55 is provided as a latching ferrite circulator, aninsulated wire routed through a ferrite core carries a control signalprovided as a DC current fed through the wire and establishes magneticfields in the latching ferrite isolator 55 thereby biasing the ferriteand determining which of the circulator signal paths are provided havinga relatively low insertion loss characteristic and which circulatorsignal paths are provided having a relatively high insertion losscharacteristic (isolation). In this manner, predetermined ports of thelatching ferrite isolator are said to be through ports (i.e. signalspropagating in a particular direction from one port to the otherexperience a relatively small amount of signal attenuation) andpredetermined ports of the latching ferrite isolator are said to beisolated ports (i.e. signals propagating in a particular direction fromone port to the other experience a relatively large amount of signalattenuation).

Once the magnetic fields are established in the switch assembly 55, theyare self-sustaining (i.e. latching) and the DC current can be removed.When it is desired to change the insertion loss characteristics betweenthe ports of the switch assembly 55, a DC current having a minimumpredetermined current level is applied to offset the bias fieldestablished via the previously applied signal and to reestablish themagnetic fields in the ferrite core and thereby change levels ofinsertion loss between the various ports in a second predeterminedmanner.

Thus, in this particular application, the circulator 55 is used toselect a signal path having a relatively low insertion losscharacteristic between the antenna 51 and one of the receiver channelsor a high power RF termination (such as termination 22 in FIG. 1) duringthe receive and transmit functions, respectively. In a preferredembodiment, the switch assembly 55 provides either an isolation signalpath having an isolation characteristic of 20 decibel (dB) or more or arelatively low loss signal path having an insertion loss characteristicof about 0.5 dB.

The switch assembly 55 also provides a convenient location for injectionof a pre-monopulse RF calibration signal. Specifically, the switch porthaving the termination coupled thereto (e.g. port 19 c in FIG. 1) can beused as a calibration signal input port as described above inconjunction with FIG. 1A.

If high power protection is not required for a particular application,switch assembly 55 may be omitted from the receiver 50 thereby reducingthe size of the receiver 50. Also, since the switch 55 introduces anadditional insertion loss to signals propagating from the antenna 51 tothe receiver circuit 56, removal or omission of the switch assembly 55also improves the noise figure of the overall system 50.

The switch assembly 55 is disposed over an RF module cover 58. Modulecover 58 has an aperture 58 a and a plurality of mounting holes 59provided therein which mate with mounting holes 59 provided in theswitch assembly 55. RF module cover 58 also includes one or morealignment pins 59 a which project into alignment holes of the switchassembly 55 to thus align the RF module cover 58 to the switch assembly55.

Disposed in the RF module cover aperture 58 a is an RF insulation cover60. The RF insulation cover 60 includes a plurality of waveguide signalpaths 62 which align with the waveguide signal paths 57 provided in boththe base portion 53 of the corrugated horn 51 and the switch assembly55. The RF insulation cover 60 also includes one or more slots 63 whichare used in a manner to be described below to insure proper alignmentbetween the waveguides 57 of the horn 51, the switch assembly 55 andwaveguides 62 in the insulation cover 60.

The receiver 50 further includes an RF housing 64 having a pair ofalignment bosses 66 projecting from an internal bottom surface thereof.Alignment bosses 66 engage corresponding slots 63 in the RF insulationcover 60 thereby assuring proper orientation and positioning of the RFinsulation cover 60 with respect to the waveguide 57 in the corrugatedhorn 51 and the switch 55. The alignment bosses 66 also ensure correctorientation and positioning of the waveguide signal paths 62 withrespect to mixer circuits formed on substrates 68 when the substratesare disposed in RF housing 64 and aid in the isolation between adjacentreceiver channels.

Referring briefly to FIGS. 2C, 2D, prior to disposing the RF modulecover 58 over the RF housing 64, a plurality of mixer substrates 68 aredisposed in appropriately shaped recessed portions 64 a (FIG. 2D)provided in an internal base portion of the RF housing 64 only a portionof which is shown in FIG. 2D. Disposed on the substrate 68 is a mixercircuit 69 (FIG. 2D). Each of the mixer substrates 68 includes aprojecting portion 70 (FIG. 2D) which form so-called E-plane probeswhich couple electric field energy propagating substantially in thewaveguide 62 in the TE10 waveguide mode into a microstrip mode to thusefficiently transition the W-band waveguide signals in the RF insulationcover 60 to subharmonic mixers 69 (FIG. 2D) in each receiver channel.Thus, the projecting portions 70 of the mixer substrates and theassociated transmission line disposed thereon formwaveguide-to-microstrip transition circuits.

Mixer diodes 71 and transmission lines 72 are disposed on the mixersubstrate 68 to thus form the mixer circuit 69. The performance of thewaveguide-to-microstrip transition circuit can be optimized via awaveguide backshort assembly 73 formed by spacing a bottom surface ofthe recess region 64 a in the area of projecting region 70 apredetermined distance from the E-plane probe 70 a disposed onprojecting region 70. Portions of the substrate 70 have here beenremoved to expose the back short 73. The electric field configurationestablished by backshort 73 can be adjusted via the use of shims orother appropriate mechanism to allow the electric field signal to beefficiently coupled to the E-plane probe 70 a. Ideally, such a backshortassembly establishes a short circuit impedance at particular physicallocation in the waveguide. In a preferred embodiment, the backshortassembly is disposed at a distance corresponding to an odd multiple of aquarter wave length (in the waveguide medium) from the protruding probe70 a.

Referring again to FIGS. 2 and 2C, also disposed in the RF housing 64are LO and IF distribution circuit substrate 75 and an IF phasingsubstrate 76. The LO and IF distribution substrate 75 includes LO and IFdistribution circuits to distribute the local oscillator andintermediate frequency signals to appropriate ports in the RF housing64. The IF phasing substrate 76 is used to provide coarse phasecompensation to phase match each of the receiver channels and providesadditional IF signal amplification.

Once the mixer substrates 68, LO and IF distribution substrate 75, andIF phasing substrate 76 are disposed in housing 64, the RF insulationcover 60 is disposed thereover. The RF module cover 58, switch 55 andhorn 51 can then be coupled to the RF module housing 64.

The RF housing 64 also has provided therein a plurality of openings 88through which corresponding ones of IF interface connectors 90 aredisposed to form an RF connection between the RF housing 64 and an IFhousing 92. The IF housing 92 has an IF substrate 94 disposed therein.Disposed on IF substrate 94 is a monolithic microwave integrated circuit(MMIC) monopulse comparator network. Suffice it here to say that theMMIC monopulse comparator network receives signals fed thereto and formsmonopulse output signals at output ports 92 a-92 d of the IF housing 92.An IF module cover 96 is disposed over the open surface of IF housing 92and an IF evacuation housing 98 is disposed over a second oppositesurface of the IF housing 92. The IF assembly is evacuated via anevacuation port 98 a provided in the IF evacuation housing 98. The IFmodule can also include amplitude and phase adjustment circuits similarto amplitude and phase adjustment circuits 36 described above inconjunction with FIGS. 1 and 1A.

The mixer circuits 69 feed IF signals to corresponding ones of LNAs 30(FIG. 2C) which in turn provide amplified IF signals via feedthroughcircuits 90 to amplitude and phase adjustment circuits 36 (FIG. 2C) andmonopulse circuit 40 (FIG. 2C).

As can be seen in FIG. 2, the RF housing 64 has projecting from a bottomsurface thereof a pair of cryostat tube housings 80. The cryostat tubehousings 80 accept cryostat assemblies 82 used to cool the mixer diodesand the low noise amplifiers thereby reducing the noise figure of thereceiver assembly 56. An RF evacuation housing 84 is disposed over thecryostat tube housings and cryostat assemblies 82 and coupled to thebottom surface of the RF housing 64 via an air tight seal formed by agasket 86. The RF assembly is evacuated via an evacuation port 86 aprovided in the RF evacuation housing 84.

In one embodiment, the cryogenic cooling hardware is fabricated fromstainless steel. Two cryostats deliver Argon gas to the back surface ofthe RF housing 64, beneath the mixer diodes 71 (FIG. 2D). The twocryostats are disposed inside a dewar assembly which also functions asthe evacuation chamber. A partial vacuum is pulled throughout thedewar/receiver assembly all the way to the sealed horn assembly. Vacuumseals are maintained at all mechanical interfaces between the horn anddewar assembly with greased O-rings.

Typical operating characteristics of a fully integrated W-Band receiver50 including horn and switch assemblies are shown in Table 1. Thisreceiver performance was determined assuming enhancement due to coolingof the mixer diodes.

TABLE 1 Parameter Performance RF to IF Conversion Gain GT 20 dB NoiseFigure LT 8 dB Output I dB CP NLT + 8 dBm Output TOI NLT + 18 dBm DeltaChannel Null Depth LT 25 dB

As also shown in FIG. 2, the receiver 50 has been partitioned into twomain housings. In this embodiment, the RF housing 66 captures the RFenergy, splits the captured RF signal into four channels anddownconverts the RF signal to the C-Band frequency range. The C-Bandfrequency signals are fed to the low noise amplifiers (LNAs) 30 in theRF module which provide low noise amplification. The amplified C-bandfrequency signals are coupled from output ports 88 of the RF housing toinput ports of the IF housing 92 via the connectors 90. The IF housing92 includes phase and amplitude adjustment circuits similar to circuits36 described above in conjunction with FIG. 1 which are used toappropriately adjust the amplitude and phase of the C-band frequencysignals before the four signals are processed by a monopulse circuitwhich generates monopulse C-Band output signals of the receiver 50

The cryogenically cooled, low noise figure, millimeter wave, monopulsereceiver 50 can be configured to adapt to a quasi-optical transmissionline which forms the feed for a large diameter reflector antenna. Such aunit may also be used independent of the reflector to provide wider beammonopulse antenna performance for small projectiles or be placed in thefocal plane of a lens assembly to provide appropriate feed illuminationcharacteristics. In addition to missile radar applications, otherconfigurations of the basic mixer architecture (cooled and uncooled)could be useful for various other commercial millimeter waveapplications such as: weather radars, space based radiometers andsatellite and airborne imaging systems.

Referring now to FIG. 3 an RF receiver assembly 100 is shown fabricatedas a single module millimeter wave integrated circuit (MIC) having alocal oscillator signal input port 100 a, a calibration signal inputport 100 b, IF monopulse signal output ports 88 a-88 d and a pair of DCbias terminals 100 c. The RF mixer circuit 69 is disposed on a substrate68, here comprised of 0.005 inch thick alumina or other suitablemicrowave/millimeter wave substrate material, such as quartz, fusedsilca, gallium arsenide or any other suitable material know to those ofordinary skill in the art. The substrate 68 has disposed on a bottomsurface thereof, a ground plane conductor 102. The substrate 68 hasdisposed over a top surface thereof a plurality of strip conductors 105which form RF signal paths and an anti-parallel diode pair 104.

LO signals fed to LO input port 100 a are coupled through a powerdivider 106 and fed to second and third power dividers 108, 110 whichfurther divide the power and feed the LO signal to respective ports ofmixer circuits 69 a-69 c. With the mixers properly biased, RF signalsfed via the waveguides to the anti-parallel diode pairs 104 produceintermediate frequency signals at mixer IF output ports which arecoupled to input ports of monolithic microwave integrated circuit (MMIC)low noise amplifiers (LNA's) 112. Each of the LNA's 112 provide the IFsignals fed thereto to a corresponding one of a plurality of input portsof a MMIC monopulse network comparator 116. The MMIC monopulsecomparator forms monopulse signals and provides the monopulse signals toIF output ports 88 a-88 d.

It should be noted that in this particular embodiment, the receiver 100is disposed in a single housing 101 (in contrast to the multiple housingapproach described above in conjunction with FIG. 2).

In operation, W-Band signals enter the RF assembly through waveguidetransmission lines, such as a WR 10 waveguide transmission line,provided in the gold-plated stainless steel cover 60 (FIG. 2). An LOsignal in the Q-band frequency range is fed through an RF connector suchas a 2.4 millimeter (mm) coaxial connector at the input port 100 a. TheLO signal is coupled through a series of Wilkinson-type power splitters106, 108, 110 and fed to each of the mixer circuits 69 a-69 d.

Each of the mixers 69 a-69 d provides an IF signal to an input of agallium-arsenide (GaAs) MMIC LNA 112 which amplifies each of the IFoutput signals. The amplified IF output signals are coupled from the LNA112 to respective input ports of a monopulse comparator circuit 116. Themonopulse circuit 116 receives the IF signals fed thereto and generatesmonopulse output signals at IF output ports 88 a-88 d.

In one particular embodiment, the monopulse comparator circuit isfabricated using GaAs MMIC processing techniques and forms the sum anddifference signals using MMIC lumped circuit elements.

The IF output signals and the calibration signal are coupled to therespective ports 88 a-88 d and 100 b through OSMP glass bead coaxialconnectors. Also coupled to housing 101 are a pair of reference voltagebias terminals 100 b, 100 c through which a DC bias signal is coupled tothe two pairs of low noise amplifiers. In one particular embodiment, thebias source is provided as a +5 V DC bias source and the LNAs areselected to draw approximately 80 mA of current.

The waveguide-to-microstrip probe transition 70 feeds each W-Band signalinto a diode pair of a subharmonic mixer. The diode pairs may beprovided, for example, as back-to-back Schottky diodes responsive tosignals in the frequency range of interest. The diodes mix the W-Bandsignal with the second harmonic of the Q-Band Local Oscillator input togenerate an IF signal having a frequency in the C-Band frequency range.

Each of the Alumina mixer substrates 68 are provided having a thicknesstypically of about 0.005 inch (5 mils) to ensure proper mixer operationin the W-Band frequency range. The substrate, on which the LOdistribution network and all IF circuitry is disposed, is provided as anAlumina substrate having a thickness of about 0.010 inch (i.e. 10 mils).The RF assembly 100 uses a combination of epoxies and solders to achievegood adhesion and ground contact between the substrates and the housing101 for both room ambient and cryogenic temperature operation.

Referring now to FIG. 4, a partial exploded view of a receiver 120reveals a Dewar 122 having a cryostat assembly 123 coupled to a firstend there of and an RF module 125 coupled to a second end thereof. Thecryostat assembly 123 includes a cryostat base plate 133 havingcryostats 134 a, 134 b and cryostat exhaust portions 136 a, 136 bprojecting therefrom. RF module 125 includes an RF module cover 124, anRF module base portion 126 and stainless steel tubes 128 a, 128 b. Whenthe RF module 125 is coupled to Dewar 122, the module 125 is disposedover an O-ring groove 130 and tubes 128 a, 128 b accept portions ofcryostat assembly 123. RF module 125 is provided from a material havinga coefficient of thermal expansion which is relatively close to thecoefficient of thermal expansion of the substrates disposed within theRF module base 126. For example, if the circuits disposed within thebase 126 are provided as thin film Alumina microstrip circuits, the base126 may be provided from a material such as Alloy 46 which has acoefficient of thermal expansion which is relatively close to thecoefficient of thermal expansion of the thin film alumina microstripsubstrates. Those of ordinary skill in the art will appreciate of coursethat other material combinations may also be used to provided thehousing and the substrates disposed therein.

As mentioned above, in one embodiment, sub-harmonically pumped mixersuse the second harmonic of the LO signal to efficiently downconvert theW-Band RF energy incident on the four channel receiver to the desired IFfrequency where it can more easily be processed. GaAs Schottkyanti-parallel beam lead diodes are the non-linear devices used in themixer circuits 69 a-69 d. The diodes are mounted on five (5) mil thickalumina substrates where the filtering and transmission line structuresfor proper impedance matching to the diodes at the appropriatefrequencies can be etched using thin film technology.

FIG. 5 shows an exemplary implementation of a passive MMIC monopulsecomparator 200 in accordance with the present invention. In general, themonopulse 200 includes a series of lumped element hybrid circuits 202a-d coupled together so as to provide a completely passive device. Eachof the lumped element hybrid circuits 202 includes so-called pi and Tfilter circuits that together provide the required signal phase shiftsfor the monopulse circuit, as described in detail below. Thisarrangement provides dramatic space and power savings over conventionalmonopulse circuits having distributed element hybrid circuits.

The monopulse comparator 200 receives on respective input ports 204 a-dinput signals A, B, C and D, which can correspond to the downconvertedsignals provided on input ports 40 a-d from the amplitude and phaseadjustment circuits 36 in FIG. 1. From the input signals A,B,C,D, themonopulse comparator 200 provides respective sum (SUM), azimuth (ΔAZ),elevation (ΔEL), and delta-delta or Q (ΔQ) signals on output ports 206a-d for subsequent target tracking processing using techniques wellknown to one of ordinary skill in the art.

FIGS. 6-7 schematically illustrate the lumped element hybrid circuits202 of FIG. 5. In one embodiment, the hybrid circuit 202 includes one Tcircuit 250 and three pi circuits 252 a-c. The T circuit 250 includesone inductor 254 coupled between first and second capacitors 256,258.That is, the T-circuit 250 has a series capacitor-shunt inductor-seriescapacitor arrangement. The T-circuit 250 can be considered a high passfilter that replaces three-quarter wavelength lines used in conventional180 degree hybrid circuits.

The pi circuits 252 a-c have a shunt capacitor-series inductor-shuntcapacitor arrangement that provides a low pass filter circuit forreplacing one-quarter wavelength lines in known hybrid circuits. The picircuit provides a 90 degree phase shift and the T circuit provides a270 degree phase shift for a relative phase shift of 180 degrees. Thelow and high pass filter structures of the hybrid circuit 200 replacecomponents in conventional distributed element hybrid circuits, whichrely upon transmission line lengths to provide the desired signal phaseshifts. Thus, the filter structure of the passive device 200 achievessignificant space savings over transmission line based circuits,particularly for embodiments in which the passive monopulse receivesdown-converted signals. In addition, the filter structure of the presentinvention is readily scalable for use at other frequencies, unliketransmission line based circuits.

In one embodiment, the relative impedance characteristic for each of theinductors is the same, i.e., L. The relative impedance values for thecapacitors are shown as C and 2C. The capacitors furthest from the Tcircuit 250 can have a relative capacitance 2C twice the capacitance Cof the other capacitors. This results from combining shunt capacitorsfrom adjacent low pass filter (pi) circuits at the ports locatedopposite the T circuit 250.

It is understood that the impedance values for the inductors andcapacitors can vary to achieve the requisite phase shifts from lumpedelement hybrid to lumped element hybrid. One of ordinary skill in theart can readily determine optimal impedance values for the inductive andcapacitive elements of the lumped element hybrids.

In one embodiment, parasitic capacitances associated with the inductorswere determined and taken into account in selecting the capacitancevalue C of the capacitors. For example, if an inductor has an associatedparasitic capacitance of 1 pF and the desired capacitance value C forthe capacitors is 3 pF, then circuit is implemented with a nominalcapacitor value of 2 pF (3pF−1 pF). The parasitic capacitance value ofthe inductor provides, in combination with the nominal value provide thedesired operational value of 3pF. Thus, the capacitive parasitics fromthe inductors are absorbed into adjacent capacitors. The inductorquality is improved in a sense to provide better monopulse performance.

FIG. 8 schematically shows the MMIC monopulse comparator 200 of FIG. 5,which includes four lumped element hybrid circuits 202 a-d, such asthose shown in FIGS. 6-7. The relative orientation of each lumpedelement hybrid 202 is indicated with a dot in a corner of the hybrid.

The first hybrid 202 a receives input signals A and B and provides afirst output signal 1HO1 to the third hybrid 202 c and a second outputsignal 1HO2 to the fourth hybrid 202 d. The second hybrid 202 b receivesinput signals C and D and provides a first output signal 2HO1 to thethird hybrid 202 c and a second output signal 2HO2 to the fourth hybrid202 d. The third hybrid 202 c, which receives signals 1HO1,2HO1 from thefirst and second hybrids 202 a,b, provides the SUM channel signal andthe elevation (EL) channel signal. The fourth hybrid 202 d, whichreceives input signals 1HO2,2HO2 from the first and second hybrids 202a,b, provides the azimuth (AZ) and delta-delta (Q) channel signals.

In one embodiment, the filter structures (pi and T) in the hybrids 202have a characteristic impedance of about 70.7 Ohms to provide awell-matched reactive two-way power divider/combiner at each hybridinterface. In addition, the pi and T type structure duality reducesphase pushing and pulling of the filter response due to voltage standingwave ratio (VSWR) interactions. The duality of the pi and T filters alsoprovides return losses with respect to each other that are similar inmagnitude and phase response. Since the first and last element in eachfilter network are capacitors, the return losses are capacitive inphase. The filter duality also forces the insertion loss through each RFpath to have similar values over a relatively broad frequency range. Itis understood that the insertion loss is different for each network typedue to phase lead or lag associated with the orientation of the hybrid.That is, a series inductor provides a phase lag while a shunted inductorprovides a phase lead.

Connections within the hybrids 202 and connections external to themonopulse can be formed from high impedance (narrow) lines, e.g., 100Ohms, to reduce junction effects that can distort the signals.Transmission line lengths interconnecting the hybrids can have lowimpedance (wide) lines corresponding to about 90 degrees, i.e., quarterphase, to reduce VSWR interactions between the hyrbids. Otherwise,reflected signals can cancel each other due to the 180 degree round tripphase shift. The relative wideness and narrowness of the internal andexternal hybrid connection is apparent from FIG. 5.

In one embodiment, air bridges used in prior art devices were replacedwith wire-bonding from adjacent circuitry in the next higher level ofassembly to the chip I/O pads over transmission lines. This arrangementimproves channel-to-channel isolation on the passive monopulse chip toprevent a leakage signal on the order of 25 to 30 dB from “swamping out”the low amplitude delta nulls.

A C-band monopulse comparator circuit like that shown in FIG. 5 wasfabricated and tested. The SUM channel insertion loss and delta channelnull depth for each of the three delta outputs (AZ,EL,Q) were measuredvs. frequency for the case where the target was “on boresight.” As knownto one of ordinary skill in the art, ideal monopulse patterns woulddisplay 0 dB sum channel insertion loss and an infinite null depth foreach of the AZ, EL, and Q channel outputs. Results at the centerfrequency of operation are provided in Table 1 below.

TABLE 1 Parameter Result (dB) Sum Channel Insertion Loss 2.9 AzimuthChannel Null Depth 27.2 Elevation Channel Null Depth 26.9 Q Channel NullDepth 33.2

It should be noted that approximately 1.5 dB of test fixture insertionloss is included in the data presented in Table 1, so that actual sumchannel insertion loss is on the order of 1.4 dB.

FIG. 9 graphically illustrates the SUM, AZ, EL, and Q insertion lossversus frequency of the C-band device described above. As can be seenfrom the insertion loss data, a relatively broad fractional bandwidth ofapproximately 37.5 ((7−4.75)/6)(100) percent is achieved. As know to oneof ordinary skill in the art, fractional bandwidth is defined as(f_(H)−f_(L))/f_(C), where f_(H) and f_(L) are the highest and lowestfrequency respectively allowing a predetermined null depth threshold,e.g., 25 dB, for the delta channels about a center frequency f_(C).

Passive MMIC monopulse comparators in accordance with the presentinvention provide significant advantages over prior art monopulsecomparators. For example, the passive monopulse comparator of thepresent invention eliminates DC power consumption and thermaldissipation. Due to its passive nature, the passive monopulse behaveswell over a relatively wide temperature range unlike active devices.This feature of the invention can provide an important advantage overconventional active monopulse circuits, particularly in cryogenicallycooled systems.

Since similar values are used for the inductors in each orientation(shunt or series), the hybrid of the present invention is inherentlyinsensitive to processing variations. More particularly, if a seriesinductor value in the low pass filter structure (pi) is different thandesired due to processing tolerances, than the shunt inductor value willvary in a similar manner. These variations are removed in the deltaoutput where the difference operation is performed, which causes gooddelta channel null depths to be attained that are unaffected byprocessing variations. Thus, the passive monopulse of the presentinvention does not require tuning and is readily scalable for operationat a desired frequency.

Unlike standard distributed rat race hybrids, the geometric shape of thepassive monopulse is symmetric such that one input or output is locatedon each of four sides. This arrangement forces each signal to propagatethrough the same number of mitered corners on each hybrid, so that anyparasitic loss or phase shift incurred by a signal passing through acorner is similarly incurred by the other monopulse signals.

The passive monopulse of the present invention also providesconsiderable space savings over known transmission line-based monopulsehybrids. For example, a C-band passive monopulse in accordance with thepresent invention occupies a total volume of about 0.00012 cubic inches(0.161″×0.190″×0.004″). A frequency scaled L-band version measures about0.00018 cubic inches (0.187″×0.239×0.004″). An increase in wavelength ofabout 300 percent requires a concomitant chip size increase of about 50percent. A conventional C-band monopulse can be about 13,500 timeslarger than a passive C-band monopulse in accordance with the presentinvention.

Accordingly, having described preferred embodiments of the invention, itwill now become apparent to one of ordinary skill in the art that otherembodiments incorporating their concepts may be used. It is felttherefore that these embodiments should not be limited to disclosedembodiments, but rather should be limited only by the spirit and scopeof the appended claims. All references cited herein are incorporatedherein by reference in their entirety.

What is claimed is:
 1. A passive MMIC monopulse comparator circuitcomprising: a first lumped element hybrid circuit; and a second lumpedelement hybrid circuit coupled to the first lumped element hybridcircuit, wherein the first lumped element hybrid circuit and the secondlumped element hybrid circuit form a part of the passive MMIC monopulsecomparator circuit.
 2. The circuit according to claim 1, wherein each ofthe first and the second humped element hybrid circuits includes a Tcircuit coupled to a pi circuit.
 3. The circuit according to claim 2wherein the T circuit is a series capacitor, shunt inductor, and seriescapacitor type T circuit.
 4. The circuit according to claim 2, whereinthe pi circuit is a shunt capacitor, series inductor, shunt capacitortype pi circuit.
 5. The circuit according to claim 1, wherein at leastone of the first and the second lumped element hybrid circuits includesa first series circuit path including a first capacitor, a firstinductor and a second capacitor, a second series circuit path includinga third capacitor, a second inductor and a fourth capacitor, a thirdseries circuit path including fifth and sixth capacitors, a fourthseries circuit path including the first and second inductors, a thirdinductor and the fifth and sixth capacitors, and a fourth inductorcoupled between the fifth and sixth capacitors.
 6. The circuit accordingto claim 5, wherein the first and third capacitors have a capacitancevalue that is about double a capacitance value of the second, four,fifth and sixth capacitors.
 7. A passive MMIC monopulse comparator,comprising: a first lumped element hybrid circuit receiving first andsecond input signals and providing first and second output signals; asecond lumped element hybrid circuit receiving third and fourth inputsignals and providing third and fourth output signals; a third lumpedelement hybrid circuit receiving the first output signal from the firstlumped element hybrid circuit and the third output signal from thesecond lumped element hybrid circuit and providing a sum channel outputsignal and an elevation channel output signal; and a fourth lumpedelement hybrid circuit receiving the second output signal from the firstlumped element hybrid circuit and the fourth output signal from thesecond lumped element hybrid circuit and providing an azimuth channeloutput signal and a Q channel output signal.
 8. The monopulse comparatoraccording to claim 7, wherein the first lumped element hybrid circuitincludes a plurality of pi-type circuits and a T type circuit.
 9. Themonopulse comparator according to claim 7, wherein the first lumpedelement hybrid circuit includes a first series circuit path including afirst capacitor, a first inductor and a second capacitor, a secondseries circuit path including a third capacitor, a second inductor and afourth capacitor, a third series circuit path including fifth and sixthcapacitors, a fourth series circuit path including the first and secondinductors, a third inductor and the fifth and sixth capacitors, and afourth inductor coupled between the fifth and sixth capacitors.
 10. Themonopulse comparator circuit according to claim 9, wherein the fifth andsixth capacitors and the fourth inductor comprise a T type circuit. 11.The monopulse comparator circuit according to claim 9, wherein the firstcapacitor, the first inductor, and the second capacitor comprise a pitype circuit.
 12. A radar system, comprising: an antenna having aplurality of antenna ports, the antenna responsive to signals having afrequency in a predetermined frequency range; a plurality of protectioncircuits each of the protection circuits having a first port coupleddirectly to a respective one of the plurality of antenna ports and asecond port, each protection circuit responsive to a first controlsignal which allows signals to propagate in a first direction betweenthe first port of the protection circuit and the second port of theprotection circuit and each protection circuit also responsive to asecond control signal which allows signals to propagate in a secondopposite direction between the first port of the protection circuit andthe second port of the protection circuit; a receiver circuit fordownconverting signals from the second ports of the protection circuitsto IF signals, the receiving circuit including a passive MMIC monopulsecomparator.
 13. The system according to claim 12, wherein the receivingcircuit further includes: a plurality of mixers, each of the pluralityof mixers having a first port coupled to a respective one of theplurality of protection circuits, a second port for receiving a biassignal, and a third port for providing a frequency shifted signal; and aplurality of amplifiers, each of the plurality of amplifiers having afirst port coupled to the third port of a respective one of theplurality of mixers and a second port coupled to the monopulsecomparator.